1. Field of the Invention
The present invention relates to a wireless communication terminal for performing communication in a system for mobile communication, satellite communication or mobile satellite communication. More specifically, the present invention relates to a frequency error estimator in a demodulator for recovering a received signal.
2. Description of the Background Art
Following the recent development of wireless communication, mobile communication, satellite communication and mobile satellite communication are widely utilized. Such communication is performed through a communication system such as an FDMA (frequency division multiple access) system or a TDMA (time division multiple access) system. In such a transmission system, data are transmitted in units referred to as bursts. In each of an SCPC (single carrier per channel) system in the FDMA system and the TDMA system, an unmodulated part for recovering a carrier, for example, is transmitted for correcting frequency deviation of a transmitted signal resulting from a Doppler effect or the like and performing synchronized recovering.
FIG. 1 illustrates an exemplary structure of a burst employed for data transmission. Referring to FIG. 1, the burst includes a carrier recovering code CR (carrier recovery), a clock recovering code STR (symbol timing recovery), a unique word UW and a data part DATA. A silent part by voice activation in the SCPC system or a silent part GT of a guard time in the TDMA system is arranged outside this burst.
The data part DATA may include a synchronizing control signal and a line control signal.
The carrier recovering code CR, which is formed by an unmodulated carrier, has a pattern xe2x80x9c11xe2x80x9d or xe2x80x9c00xe2x80x9d. The clock recovering code STR is obtained by modulating a carrier in accordance with a clock signal. The unique word UW, which is an identification code indicating that the burst starts from this position, is a modulation code having a known pattern. The data part DATA, the structure of which varies with the structure (reference burst/data burst) of this burst and a line connection system, includes at least data necessary for line connection.
Carrier recovering is performed on a received wave of a burst form using the carrier recovering code CR which is an unmodulated signal, and clock recovering is performed with the clock recovering code STR for recovering the received signal.
Communication data is transmitted while modulating a carrier with the data. A receiver must demodulate the received modulated signal. The demodulation system includes an FSK (frequency shift keying) system, a PSK (phase shift keying) system and the like. Consider a QPSK system as a representative demodulation system, in order to simplify the description.
Transmission data are divided into pairs of bits to be transmitted. A 2-bit pair represents one of four states (11), (01), (00) and (10). These four states (11), (01), (00) and (10) are assigned to phases xcfx80/4, 3xcfx80/4, 5xcfx80/4 and 7xcfx80/4 and modulated. In this QPSK system, one of two bits of data is multiplied by a carrier and the other is multiplied by a signal obtained by phase-shifting the carrier by xcfx80/2. These signals are referred to as an I-channel signal and a Q-channel signal, respectively. The I-channel signal and the Q-channel signal are added up and then transmitted. In relation to this QPSK system, consider a xcfx80/4 shift QPSK system having eight modulated states and limiting modulated states for subsequent transition to xc2x1xcfx80/4 and xc2x13xcfx80/4.
FIG. 2 shows the positions of the modulated states of the xcfx80/4 shift QPSK system. Referring to FIG. 2, states P0 to P7 are arranged out of phase by xcfx80/4 from each other. State transition is limited to xc2x1xcfx80/4 and xc2x13xcfx80/4, and hence the state P0 can be shifted to one of only the four states P1, P3, P5 and P7, for example. In this xcfx80/4 shift QPSK system, band limitation is simple since state transition is made without passing the origin while no excessive sideband is caused since fluctuation of the envelope is small, whereby communication of a number of stations can be made at channel intervals.
In this wireless communication terminal, it is necessary to estimate a frequency error caused by drift of a local oscillator or the like for eliminating this frequency error and sampling received data thereby correctly recovering the received data. Therefore, a demodulator is provided with a frequency error estimator. In relation to such a frequency error estimator, a single-symbol delay detection frequency error estimator and a multi-symbol delay detection frequency error estimator are proposed. In TECHNICAL REPORT OF IEICE, SANE 95-114, SAT95-92 (February 1996), for example, Kubo et al. describe a multiple circuit AFC as a multi-symbol delay detection frequency error estimator in an article titled xe2x80x9cA Multiple Open-Loop AFC for MPSKxe2x80x9d.
FIG. 3 schematically illustrates the structure of a conventional single-symbol delay detection frequency error estimator. Referring to FIG. 3, the conventional one-symbol delay detection frequency error estimator includes a phase converter 13 receiving an I-channel signal supplied to an input terminal 11 and a Q-channel signal supplied to an input terminal 12 and detecting the phase of a received QPSK modulated signal from the I-channel signal and the Q-channel signal, a single-symbol delay detector 15 obtaining the difference between phase information output from the phase converter 13 and phase information preceding by one symbol and extracting phase difference information, an orthogonal transformer 16 generating I-channel data and Q-channel data in accordance with the phase difference information from the single-symbol delay detector 15, moving average filters 17 and 18 obtaining moving averages of the I-channel data and the Q-channel data from the orthogonal transformer 16 respectively, a power calculator 19 calculating the power of the received signal in accordance with output components from moving average filters 17 and 18, a comparator 20 comparing the power output from power calculator 19 with a predetermined threshold, and a frequency error calculator 21 activated in response to activation of an unmodulated signal detection signal CRDT from comparator 20 for generating a frequency error, EFER from the I-channel component and the Q-channel component from moving average filters 17 and 18.
The unmodulated signal detection signal CRDT from comparator 20 is supplied to a synchronization establish circuit through an output terminal 23, while the estimated frequency error signal EFER from frequency error calculator 22 is supplied to a voltage control oscillator (VCO) generating a recovered carrier signal through a terminal 22. Operations of the frequency error estimator shown in FIG. 3 are now described.
In order to simplify the description, it is assumed that the received signal (IQ signal) is a signal demodulated in accordance with the xcfx80/4 shift QPSK system. General description is provided later. In this case, two-bit symbol data are differentially encoded. In this differential encoding, the following phase information is differentially encoded and transmitted:
xcfx86xe2x80x2(n)=xcfx86xe2x80x2(nxe2x88x921)+xcfx86(n)xe2x80x83xe2x80x83(1),
where xcfx86xe2x80x2(n) and xcfx86xe2x80x2(nxe2x88x921) represent the phase of a currently transmitted symbol, the phase of a signal transmitted precedently by one symbol, and xcfx86(n) represents a phase corresponding to the data to be transmitted. In the xcfx80/4 shift QPSK system, the phase xcfx86(n) is either one of xc2x1xcfx80/4 and xc2x13xcfx80/4.
Phase converter 13 receives an I-channel component and a Q-channel component supplied to input terminals 11 and 12 respectively and detects the their phases. A received signal Sd(t) is expressed as follows:                               Sd          ⁡                      (            t            )                          =                  I          +                      j            ·            Q                                                            =                      ext            ⁢                          {                              j                ⁡                                  (                                                            2                      ⁢                                              xe2x80x83                                            ⁢                                              π                        ·                        fc                        ·                        t                                                              +                                          φ                      xe2x80x2                                                        )                                            }                                      ,            
where fc represents the frequency of the carrier and xcfx86xe2x80x2 represents the phase of the received IQ signal.
Q/I=tanxe2x88x921(2xcfx80xc2x7fcxc2x7t+xcfx86xe2x80x2)
The difference between the current symbol and the previous symbol is obtained in transmission. When setting a symbol rate and a carrier frequency so that fcxc2x7t is a multiple of an integer in the above equation, therefore, the above equation is reduced as follows:
Q/I=tanxe2x88x921(xcfx86xe2x80x2).
Hence, the phase (phase difference) xcfx86xe2x80x2 is obtained by obtaining the inverse tangent (arc tangent) of the component values of the I-channel signal and the Q-channel signal by phase converter 13.
Single-symbol delay detector 15 includes a single-symbol delayer 15a delaying the phase information from phase converter 13 by one symbol period and a subtracter (adder) 15b subtracting phase information output from single-symbol delayer 15a from the phase information output from phase converter 13. Therefore, single-symbol delay detector 15 outputs the following phase information:
xcfx86xe2x80x2(n)xe2x88x92xcfx86xe2x80x2(nxe2x88x921)=xcfx86xe2x80x2(nxe2x88x921)+xcfx86(n)xe2x88x92xcfx86xe2x80x2(nxe2x88x921)=xcfx86(n)
Thus, the phase of the received data can be detected with the output from single-symbol delay detector 15.
In the xcfx80/4 shift QPSK system, the following equation holds:
xcfx86xe2x80x2(n)=xcfx86xe2x80x2(nxe2x88x921)+xcfx86(n)
xcfx86(n) is either one of xc2x1xcfx80/4 and xc2x13xcfx80/4.
The phases of combinations (xcfx80/4, 3xcfx80/4, xe2x88x923xcfx80/4, xe2x88x92xcfx80/4) and (0, xcfx80/2, xcfx80, xe2x88x92xcfx80/4) alternately appear in the transmission phase xcfx86(n).
Therefore, the I-channel component and the Q-channel component can be derived from this phase difference xcfx86(n).
In accordance with the phase xcfx86(n) output from single-symbol delay detector 15, the orthogonal transformer 16 detects the I-channel data and the Q-channel data. As shown in FIG. 5, the phase differences xcfx80/4, 3xcfx80/4, xe2x88x923xcfx80/4 and xe2x88x92xcfx80/4 correspond to data, respectively. The I-channel data and the Q-channel data may be calculated through utilizing the equality of the tangent of the phase difference xcfx86(n) to Q/I (by utilizing a table memory, for example).
In differential encoding of the general QPSK system, detected phase differences are 0, xc2x1xcfx80/2 and xcfx80, and each phase difference is related to a symbol when no frequency error occurs.
Moving average filters 17 and 18 obtain averages of the I-channel data and the Q-channel data from orthogonal transformer 16 over prescribed symbols and eliminate harmonic components. The moving average filters 17 and 18 execute filter processing expressed in the following equations:
xe2x80x83Ifout=xcexa3I(nxe2x88x92j)/L,
Qfout=xcexa3Q(nxe2x88x92j)/L,
where Ifout and Qfout represent the output data from moving average filters 17 and 18. The summation is performed over j=0xe2x88x92(Lxe2x88x921). The tap number (Lxe2x88x921) of moving average filters 17 and 18 is set on the basis of a symbol number L of an unmodulated signal area, i.e., the carrier recovering code CR.
A total sum calculator 19 calculates the sum of squares of the I-channel data Ifout and the Q-channel data Qfout output from moving average filters 17 and 18. The power of the signal Sd(t) is obtained by the sum of squares of the I-channel data Ifout and the Q-channel data Qfout from the relation that the transmitted signal Sd(t) is equal to I+jxc2x7Q.
The comparator 20 compares the power calculated by power calculator 19 with the predetermined threshold and determines presence/absence of an unmodulated signal on the basis of the result of comparison. Assuming that PMAX represents power obtained when the moving average filters 17 and 18 are filled with a result of delay detection of the unmodulated signal in a noiseless state, xcex1xc2x7MAX is selected as the threshold. The coefficient xcex1 represents tolerance set depending on transmission path conditions etc.
When the unmodulated signal detection signal CRDT from comparator 20 is activated and a determination is made that the unmodulated signal is received, the frequency error calculator 21 calculates an average frequency error of one symbol from the I-channel data Ifout and the Q-channel data Qfout from moving average filters 17 and 18. It is assumed that fcxc2x7t is an integer in the term of 2xc2x7xcfx80xc2x7fcxc2x7t in the aforementioned encoding. If an input QPSK modulated wave has a carrier frequency error xcex94F, therefore, the input modulated wave Sd(t) is expressed as follows:
Sd(t)=exp[j{2xcfx80xc2x7xcex94Fxc2x7t+xcfx86xe2x80x2(t)}],
where t represents a discrete time expressed as nxc2x7T and T represents the symbol cycle.
In the structure shown in FIG. 3, the phase converter 13 obtains the phase of the input modulated signal from the input modulated signals (the I-channel signal and the Q-channel signal). When the input modulated wave has the carrier frequency error xcex94F, the phase converter 13 also detects the frequency error xcex94F and this frequency error component is supplied to single-symbol delay detector 15. At this time, the following phase error results from the frequency error for the current phase component:
2xcfx80xc2x7xcex94Fxc2x7T=2xcfx80xc2x7xcex94Fxc2x7T/Fs.
The single-symbol delay detector 15 performs one-symbol delay and hence the phase component error is 2xc2x7xcfx80xcex94Fxc2x71/Fs. The orthogonal transformer 16 calculates the I-channel data and the Q-channel data inclusive of the phase error component resulting from the frequency error, and hence the I-channel component Ifout and the Q-channel component Qfout from moving average filters 17 and 18 also include errors resulting from the frequency error.
Therefore, the frequency error calculator 21 performs arc tangent operation, in order to calculate the phase information xcfx86 from the I-channel component Ifout and the Q-channel component Qfout. Thus, phase information in the unmodulated signal area is obtained.
The carrier recovering code CR, which is a unmodulated signal and the carrier itself, has the pattern (symbol) 11 or 00 and regularly remains at the same shift phase quantity, and a symbol-to-symbol phase difference there of can be regarded as zero. In this case, therefore, the phase difference, i.e., the phase information xcfx86 results from the frequency error xcex94F. Hence, the frequency error xcex94F is expressed as follows:                               Δ          ⁢                      xe2x80x83                    ⁢          F                =                              (                                          1                /                2                            ⁢                              xe2x80x83                            ⁢                              π                ·                T                                      )                    ·          φ                                        =                              (                                          Fs                /                2                            ⁢                              xe2x80x83                            ⁢              π                        )                    ·                                    tan                              -                1                                      ⁡                          (                              Qfout                /                Ifout                            )                                          
Thus, the frequency error can be obtained through the unmodulated signal. In accordance with the detected frequency error EFER, an oscillation frequency is so controlled that an oscillation signal from a local oscillator generating a reference frequency signal for sampling and recovering the input signal is synchronized with the carrier. Phase synchronization of a sampling clock signal is established with the symbol timing recovering code STR, for executing sampling and recovery of data included in the data part.
Operations of the frequency error estimator are now generally described. When the received signal is a continuous wave and frequency deviation is caused, the phase regularly rotates in one direction. Consider that a received signal (In, Qn) has a phase xcex8(n) and a next received signal component (In+1, Qn+1) has a phase difference xcex94xcex8(n+1) with respect to the received component (In, Qn) as shown in FIG. 4. In this case, the phase converter 13 first calculates the phase components xcex8(n) and xcex8(n+1) simply by arc tangent operation. The single-symbol delay detector 15 generates the following phase difference component:
xcex94xcex8(n+1)=xcex8(n+1)xe2x88x92xcex8(n).
In other words, arc tangent operation is simply performed on an input received sample signal. No data value of the received signal component is taken into consideration. In this case, the orthogonal transformer 16 performs tangent operation of the phase difference xcex94xcex8(n+1) and generates an I-channel component and a Q-channel component.
In other words, the orthogonal transformer 16 obtains cos(xcex94xcex8(n)) and sin(xcex94xcex8(n)) as the I-channel component and the Q-channel component respectively, as shown in FIG. 4.
Therefore, the moving average filters 17 and 18 perform the following calculations:
Ifout(n)=xcexa3cosxcex94xcex8(nxe2x88x92j)
Qfout(n)=xcexa3sinxcex94xcex8(nxe2x88x92j)
The summation is performed as to j=0xe2x88x92Lxe2x88x921. If nxe2x88x92j less than 0, it is assumed that xcex94xcex8(nxe2x88x92j)=0. Thus, a moving average of L stages is obtained.
The power calculator 19 obtains power P(z) of filter outputs shown in FIG. 5 from the outputs Ifout(n) and Qfout(n) of moving average filters 17 and 18 as follows:
P(z(n))=Ifout(n)2+Qfout(n)2.
When the phase difference AO is distributed only in one direction, change of the envelope is small and hence the power P(z(n)) is maximized. When a modulated component is included, the direction of change thereof varies with data as described above with reference to the xcfx80/4 shift QPSK system, and hence the phase difference xcex94xcex8 change in positive and negative directions. In this case, therefore, the envelope remarkably changes and the power P(z(n)) is reduced.
The frequency error calculator 21 calculates the average frequency error AF of one symbol in accordance with the equation xcex94xcex8=tanxe2x88x921(Qfout(n)/Ifout(n)). This phase difference xcex94xcex8 is related to the average frequency error xcex94F as follows:                               Δ          ⁢                      xe2x80x83                    ⁢          F                =                  Δ          ⁢                      xe2x80x83                    ⁢                      θ            /                          (                              2                ⁢                                  xe2x80x83                                ⁢                                  π                  ·                  T                                            )                                                              =                              (                                          Fs                /                2                            ⁢                              xe2x80x83                            ⁢              π                        )                    ·                                                    tan                                  -                  1                                            ⁡                              (                                  Qfout                  /                  Ifout                                )                                      .                              
Thus, the frequency error xcex94F can be calculated similarly to the above equation. In this case, however, it is assumed that the symbol-to-symbol phase difference results from the frequency error. It is assumed that the unmodulated signal component is repetition of the same pattern and the phase error results from frequency deviation.
When the unmodulated signal (carrier recovering code CR) is formed by L symbols as shown in FIG. 6, the window of the moving average filter is an L-symbol interval. Consider the power of the carrier recovering code CR (unmodulated signal part), the noise part and the data part. When the unmodulated signal is a carrier, the input signal Sd(t) is expressed as follows, as described above:
Sd(t)=exp(jxc2x72xcfx80xc2x7xcex94Fxc2x7t)+k(t),
where k(t) represents a noise component.
On the basis of the phase component 2xcfx80xc2x7xcex94Fxc2x71/Fs, the power calculator 19 obtains the sum of squares of the respective components. In this case, therefore, the power P(z) of the sum of the squares for a single input signal is higher than a certain reference value since the phase regularly rotates only in one direction in the case of the carrier.
When only the noise component is present, the input signal has only the noise component k(t). In this case, therefore, no frequency error component contributes as compared with the carrier component while the phase changes at random in positive and negative directions, and hence the square sum P(z) of the I-channel component and the Q-channel component of the noise is sufficiently smaller than the comparing reference value.
The data part is a modulated signal component, and the symbol recovering code, the unique word and the data part can be regarded as pseudo noise (PN). Also in the case of the data part, therefore, the phase changes substantially at random in the positive and negative directions similarly to the noise and hence the power thereof is sufficiently smaller than the threshold. In the data part, the received signal Sd(t) is expressed as follows:
Sd(t)=exp{j(2xc2x7xcfx80xcex94Fxc2x7t+D)}+k(t),
where D represents a random variable indicating phase modulation.
Therefore, whether or not an unmodulated signal, i.e., a carrier arrives and whether or not a burst is input can be determined by comparing the output value P(z) of power calculator 19 with the threshold.
As described above, however, each of moving average filters 17 and 18 obtains the moving average of L symbols. When moving average filters 17 and 18 store all L symbols of the unmodulated signal, i.e., the carrier recovering code CR, therefore, the output values of moving average filters 17 and 18 are maximized.
Consider that comparator 20 has thresholds A and B as shown in FIG. 8 (the threshold A is greater than the threshold B). In this case, the moving average filters 17 and 18 perform moving average processing on signal components in a modulated signal area 31 (this area may include a noise component or data of a preceding burst) preceding an unmodulated signal (carrier) (carrier recovering code CR) included in an unmodulated signal area 32. As described above, the power in the unmodulated signal area 32 is higher than that of the data and noise parts. In moving average processing in moving average filters 17 and 18, therefore, the power of the output values of moving average filters 17 and 18 increases as the ratio of unmodulated signal area 32 increases. The unmodulated signal detection signal CRDT is asserted at a time ta in the case of the threshold B while the unmodulated signal detection signal CRDT is asserted at a time tb in the case of the threshold A. Thus, the assert timing for the unmodulated signal detection signal CRDT varies with the levels of the thresholds.
Both of the thresholds A and B are rendered smaller than the power PMAX obtained through moving average processing on all L symbols included in the unmodulated signal area 32 in practice, in consideration of transmission path loss or the like. Upon estimating an average frequency error of one symbol inclusive of a modulated signal, therefore, the number of unmodulated signal components averaged in moving average filters 17 and 18 is reduced and hence estimation precision of the frequency error estimated in frequency error calculator 21 is disadvantageously deteriorated.
When the sum of the squares of the I-channel component Ifout and the Q-channel component Qfout from moving average filters 17 and 18 exceeds the prescribed threshold, the unmodulated signal is determined to be present, and the frequency error is calculated. In this case, therefore, the position of the burst cannot be specified although presence/absence of the burst can be determined and the position for starting synchronization establishment of the symbol and sampling cannot be correctly specified.
Further, detection precision for the frequency error also remarkably depends on the set value of the threshold, and hence it is difficult to correctly set the threshold in consideration of factors such as loss on the transmission path varying with the situation of usage, and the frequency error cannot be regularly correctly estimated.
The aforementioned problems arise in a demodulator employed in a communication system transmitting data not only in the modulation system such as the xcfx80/4 shift QPSK system but also in a modulation system such as the FSK system or a general QPSK system. In other words, the problems regularly arise in a demodulator for correcting deviation of a carrier frequency with an unmodulated signal (carrier) included in a burst employed in data communication.
An object of the present invention is to provide a wireless communication terminal capable of calculating a frequency error in high precision and correctly specifying the position of a burst.
Another object of the present invention is to provide a wireless communication terminal capable of correctly detecting an unmodulated signal area and specifying the position thereof regardless of the level of a threshold providing the detection reference for an unmodulated signal.
Briefly stated, the wireless communication terminal according to the present invention detects the maximum value of the power of an input signal over a predetermined time interval after the power exceeds a threshold. The wireless communication terminal also estimates a frequency error, using I and Q channel components providing the maximum value of the power. The wireless communication terminal asserts an unmodulated signal detection signal in accordance with detection of the maximum value.
It is possible to specify a position where moving average processing is performed on all symbols of an unmodulated signal and to calculate the frequency error only with the unmodulated signal symbols by detecting the position providing the maximum value after exceeding the threshold.
The foregoing and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings.